Tunable ferro-electric filter

ABSTRACT

The present invention quantifies and reduces losses in tunable bandpass filters having ferro-electric capacitors. Given a required insertion loss and the quality factor of a resonator, the geometric losses resulting from a particular topology for the ferro-electric capacitor and the metal losses are accounted for to quantify the allowable ferro-electric losses.

RELATED APPLICATION

[0001] This application claims the benefit of U.S. ProvisionalApplication No. 60/283,093, filed Apr. 11, 2001, titled Low Loss TunableFerro-Electric Device and Method of Characterization, which is herebyincorporated by reference.

BACKGROUND Description of Related Art

[0002] Filters, such as bandpass filters, have numerous applications incommunications and electronics. For example, in wireless communicationsa given frequency band must accommodate many wireless users. Toaccommodate so many users, stringent bandpass filtering requirementsmust be achieved because of the crowded frequency allocations provided.

[0003] At present, wireless handsets use fixed-tuned bandpass filters(BPFs) to meet their filtering specifications. The design of suchfilters is complicated because they must achieve the lowest possiblepassband insertion loss (I.L.) while simultaneously achieving aspecified large out-of-band rejection. As a specific example, considerfull band PCS CDMA handsets using fixed bandwidth filters. The PCStransmit (TX) band should have no more than −3.5 dB I.L. in-band (1850to 1910 MHz in the U.S.) while having at least a 38.0 dB out-of-bandrejection in the receive (RX) band (1930 to 1990 MHz range).

[0004] Further, this BPF must meet these specifications with a maximumconstraint on height. A typical height constraint in present dayhandsets, for example, is 4.0 mm or less. To meet these demandingelectrical requirements yet possess the smallest possible size andheight, high order (>2^(nd) order) fixed-tuned filters constructed fromeither individual coaxial resonator elements or monoblock structures areusually necessary. In addition, to satisfy out-of-band rejectionspecifications, a transmission zero is usually required, increasing I.L.at the band edge. Because of variations in ceramics and fabricationtolerances, vendors must individually adjust the characteristics offixed-tuned filters during their manufacture, driving costs higher.

[0005] Moreover, if more than one frequency band were to be supported(e.g., supporting the PCS bands in the U.S., Korea, and India) multiplefixed-tuned BPFs would be necessary, requiring extra switches whichintroduces additional loss. This is true, even if the power amplifierand low noise amplifier used have sufficient bandwidth to operate overthese multiple bands.

[0006] A tunable BPF would allow the use of one BPF over several bands,or of a lower order filter to cover a bandwidth wider than a requiredpassband at any particular time. To provide the tunability in a tunableBPF, a component capable of providing a variable capacitance istypically used.

[0007] Several structures are presently used to implement a variablecapacitor. For example, movable parallel plates have been used for manyyears as the tuner in home radios. However, such plates are far toobulky, noisy, and impractical for use in most modern applications.

[0008] Another alternative, the electronic varactor, is a semiconductordevice that adjusts capacitance responsive to an applied voltage.Because the varactor is typically noisy and lossy, particularly inapplications above 500 MHz, it is ineffective for high-frequency,low-loss applications where high performance is required.

[0009] Another alternative, a micro-electro-mechanical-system (MEMS) isa miniature switching device that may switch between capacitorsresponsive to an applied control signal. It, however, is costly,difficult to manufacture and of unproven reliability. In most cases, itprovides discrete tuning, in that a system must select between a finite(and small) number of fixed capacitors.

[0010] Ferroelectric tunable capacitors are another alternative that hasbeen attempted. Ferroelectric (f-e) materials are a class of materials,typically ceramic rare-earth oxides, whose prominent feature is thattheir dielectric constant (κ), and as a consequence, the electricpermittivity (∈) changes in response to an applied slowly varying (DC orlow frequency) electric field. The relationship of the dielectricconstant (κ) and the electric permittivity (∈) of a material is given asfollows:

∈=κ∈₀

[0011] where ∈₀ is the electric permittivity of a vacuum. At present,there are several hundred known materials that possess f-e properties.In a typical f-e material, one can obtain a range in κ by a factor of asmuch as approximately 3:1. The required DC voltage to generate such achange in κ depends on the dimensions of the f-e material over which aDC control voltage is applied. As a result of their variable dielectricconstant, one can make tunable capacitors using f-e materials, becausethe capacitance of a capacitor depends on the dielectric constant of thedielectric proximate the capacitor conductors. Typically, a tunable f-ecapacitor is realized as a parallel plate (overlay), interdigital (IDC),or a gap capacitor.

[0012] In known f-e variable capacitors, a layer of an appropriate f-ematerial, such as barium strontium titanate, Ba_(x)Sr_(1−x)TiO₃ (BSTO)is disposed adjacent to one or both conductors of a capacitor. Dependingupon the strength of the electric field applied to the f-e material andthe intrinsic properties of the f-e material selected, the capacitancechanges. Typically, below the Curie temperature, T_(c), of the f-e film,the f-e material is in the ferroelectric state and will exhibithysteresis in its response to a changing electric field. Above T_(c),f-e material is in the paraelectric state and will not exhibithysteresis. Thus, one generally picks an f-e material whose T_(c) islower than the expected operating temperature so as to operate in theparaelectric state, avoiding the hysteresis effects of the ferroelectricstate.

[0013] However, conventional f-e variable capacitors have proven to betoo lossy for use in insertion-loss-sensitive applications such ashandsets. Moreover, these devices often perform unpredictably,preventing optimal design, construction, and use of f-e tunable filters.

[0014] Accordingly, there is a need in the art for improved tunable f-efilters capable of providing a tuning range over a desired frequencyrange with low I.L. and high out-of-band rejections and methods fordesigning the same.

SUMMARY

[0015] Fixed tuned bandpass filters must satisfy stringent size,insertion loss and out of band rejection, among other requirements.Tunable filters would be useful in replacing fixed tuned bandpassfilters if they could meet these requirements. Lower order, or otherwisebetter, tunable filters might be used to tune over ranges requiringhigher order fixed tuned filters. Or a single tunable filter couldreplace more than one fixed tuned filter. However, tunable filtersrequire tunable components that have consistently shown themselves to beto high in insertion loss, unreliable, or possessing other prohibitivequalities.

[0016] It is desirable to provide a tunable bandpass filter that hassuperior insertion loss properties with respect to fixed-tuned bandpassfilters yet still achieves required rejection performance and satisfiesother requirements. It is therefore an object of this invention toprovide tunable bandpass filters incorporating ferro-electric materialsto tune the filters while maintaining a low insertion loss, meetingstringent out of band rejection requirements and satisfying otherrequirements. This is made possible by the advantageous design ofcapacitors and filters based on a correct recognition of the losscharacteristics of the ferro-electric materials.

[0017] Another object of the invention is to provide a methodology fordesigning tunable bandpass filters. This methodology quantifies andminimizes loss mechanisms in tunable ferro-electric capacitors to selectoptimal structures for a tunable bandpass filter incorporating tunableferro-electric capacitors.

[0018] The primary object of this process is to allow the user to designminimum loss BPF's that meet or exceed all other electrical andmechanical specifications placed on a conventional fix-tuned BPF that itreplaces. The meeting or exceeding of performance specifications iscritical if a tunable BPF is to replace a fix-tuned BPF in practicalapplications.

[0019] Proper f-e film characterization, along with optimum tunable BPFdesign procedures are mandatory if one is to achieve minimum losstunable BPF's that simultaneously meet a stringent rejectionspecification.

[0020] In accordance with one embodiment of the invention, a method isprovided for choosing a bandwidth and filter order for a tunablebandpass filter to satisfy an out-of-band rejection requirement and apassband insertion loss requirement. Given a topology for aferro-electric capacitor, the method calculates the non-ferro-electriclosses for the ferro-electric capacitor. Given a resonator having afirst quality factor for coupling to the ferro-electric capacitor, themethod determining the required ferro-electric loss of the f-e capacitorbased upon the calculated non-ferro-electric losses and the firstquality factor to achieve an insertion loss requirement for the tunablebandpass filter.

[0021] In accordance with another embodiment of the invention, a processis described by which a wide variety of f-e films can be efficiently andcorrectly characterized.

[0022] Further aspects and features of the invention are set forth inthe following description together with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

[0023]FIG. 1a is a plan view of a ferro-electric gap capacitor.

[0024]FIG. 1b is a cross-sectional view of the ferro-electric gapcapacitor of FIG. 1a taken along line A.

[0025]FIG. 2a is a plan view of a ferro-electric overlay capacitor,along with an accompanying DC blocking capacitor.

[0026]FIG. 2b is a plan view of the first metal layer in the overlaycapacitor of FIG. 2a.

[0027]FIG. 2c is a cross-sectional view of the overlay capacitor of FIG.2a taken along line B in FIG. 2a.

[0028]FIG. 3 illustrates an enlarged view of area C in FIG. 2a.

[0029]FIG. 4 is a plan view of a ferro-electric interdigital capacitor.

[0030]FIG. 5 is a schematic of a resonator coupled to a tunableferro-electric capacitor.

[0031]FIG. 6 is a schematic of a single pole tunable filter.

[0032]FIG. 7 is a planar circuit implementation of the single polefilter of FIG. 6.

[0033]FIG. 8a is a schematic of a double pole tunable filter having aferro-electric capacitor configured to compensate for frequency responsedistortions induced by tuning.

[0034]FIG. 8b is a schematic of a double pole tunable filter having atwo ferro-electric capacitors configured to compensate for frequencyresponse distortions induced by tuning.

[0035]FIG. 9 is a schematic of a divider network and direct currentvoltage source used to tune the two ferro-electric capacitors configuredto compensate for frequency response distortions induced by tuning shownin FIG. 8b.

[0036]FIG. 10 shows one implementation of the divider network shown inFIG. 9.

[0037]FIG. 11a is a plan view of the tunable filter shown in FIG. 8a.

[0038]FIG. 11b is a cross-sectional view of the tunable filter shown inFIG. 11a, taken along line D.

[0039] Use of the same reference symbols in different figures indicatessimilar or identical items.

DETAILED DESCRIPTION

[0040] In designing a tunable bandpass filter (BPF) for use andapplication in electronic signal processing systems, such as, forexample, communications systems, one must usually meet or exceed bothout-of-band rejection and pass band insertion loss (I.L.) requirementsas well as size, weight and other mechanical, environmental andelectrical requirements imposed on fix-tuned BPF's. Further, any suchdesigns targeted for high volume products must be manufacturable andrepeatable, with consistent unit-to-unit performance, requiring aminimum (and preferably no) added tuning or testing in-situ.

[0041] Thus, for a tunable BPF to be a commercially viable replacementfor a fixed-tuned BPF, its performance should exceed that of thefixed-tuned BPF it is replacing in terms of most or all of theelectrical and mechanical requirements. In demanding applications suchas wireless handsets, passband I.L. must be minimized to prevent placingan even greater burden on other components in the handset. If a tunableBPF has I.L. greater than the fixed-tuned BPF it is to replace, theadded I.L. may prove to be too great a burden on the overall systemperformance.

[0042] Many definitions of what defines a “pass band” may be used.Typically the pass band is defined by the points where the bandpassfilter response falls to 3.0 dB below the mid-band, or band-centerinsertion loss (I.L₀.). However, any fixed filter response can be usedto define the pass band. Higher order (more resonators) bandpass filtersare typically required to meet a specific out-of-band rejectionrequirement. But increasing the filter order will increase the I.L₀. Auseful basic relationship between filter order, topology and I.L₀. isgiven by the following equation: $\begin{matrix}{{I.L_{0}.} = {\left( {4.34*{Q_{l}/Q_{u}}} \right)*{\sum\limits_{i = 1}^{N}g_{i}}}} & (1)\end{matrix}$

[0043] where N is the filter order,

[0044] Q_(u) is the unloaded Q of the resonators used,

[0045] Q₁=f₀/BW (BW is the 3 dB passband and f₀ is the midbandfrequency), and

[0046] g_(i) are the filter element values for a given topology(Chebyshev vs. Butterworth).

[0047] Generally, a Chebyshev response is preferable as it gives asteeper rejection response compared to that given by a Butterworthfilter for a given filter order. Additionally, increasing the ripple ina Chebychev BPF further increases out of band rejection. As can be seenfrom equation (1), for a given filter order N, a larger passband resultsin lower I.L₀. as Q_(l) will decrease as BW increases. This lowerI.L_(o). comes at the expense of decreased selectivity. To regainselectivity, the filter order N must be increased, at the expense ofI.L₀. One of ordinary skill in the art of bandpass filter design willappreciate that equation (1) represents the best one can do for a givensystem requirement and filter order. Using a higher order filter (moreresonators of a given unloaded Q) quickly increases I.L₀., because theg_(i) values get progressively larger in magnitude, even as there aremore of them to sum (increased N). Note that equation (1) neglectsimplementation losses, which further increase I.L._(o,) especially asthe band edge is approached.

[0048] It can be seen from equation (1) that using a first or secondorder bandpass filter reduces I.L₀. At these lower orders, both thenumber (N) of the g_(i) coefficients decreases as well as the magnitudesof the g_(i). These low-order filters should be constructed fromresonators that have the lowest loss (highest Q_(u)) so as to give theminimum I.L₀. possible. The resulting 1^(st) or 2^(nd) order bandpassfilter will always have lower I.L₀. for a given resonator size and type(i.e., for a given Q_(u)) than the comparable fixed-tuned bandpassfilter design of higher order. Tunability allows the low ordernarrowband BPF to replace a wider band, fix-tuned BPF. A tunablenarrowband low order BPF can cover the entire band of interest,overcoming the limitation of having a narrow bandwidth. This assumesthat the desired channel (information) bandwidth is narrower than thetotal system bandwidth.

[0049] Tunable BPFs have the best chance of replacing fixed-tuned BPFsin those cases where the fixed-tuned BPF covers a system bandwidth thatis greater than that required for transmission or reception of a singlechannel. For example, a fixed-tuned BPF in a handset for operation inthe U.S. CDMA PCS band covers such a BW. It will be understood that thisis also true of U.S. cellular CDMA and many other standards. Thetechniques, methods and devices taught herein are applicable to manystandards besides U.S. CDMA PCS. U.S. CDMA PCS is discussed as anexample only.

[0050] In the full U.S. PCS band, 60 MHz is allocated for Tx (1850 to1910 MHz) and 60 MHz for Rx (1930 to 1990 MHz) for full band operation.The CDMA standard is a full duplex system, meaning the handset mustsimultaneously transmit and receive. To accomplish this, a duplexerfilter is needed to separate the bands and prevent interference. Whilethe PCS band is 60 MHz wide, the individual CDMA channel is only 1.25MHz wide. Current system architecture, however, forces CDMA PCS bandpassfilters and multiplexers (including duplexers) to have a BW≧60 MHz asthe system must allow for and accommodate operation of any 1.25 MHzchannel in any region of the 60 MHz band.

[0051] A tunable PCS band filter could alter this situation by meetingthe worst case rejection specifications while providing a lower orderBPF of simpler topology that occupies a smaller physical area. Such alower order filter would necessarily provide lower I.L₀. by virtue ofequation (1). In some circumstances partial band operation, covering aband less than 60 MHz, is desired. Tunable BPF's would be equallyadvantageous in these circumstances.

[0052] To effectively replace a high-order fixed-tuned BPF with alow-order tunable BPF, three factors should be considered. First, thefractional bandwidth (i.e. Q₁) of the low-order BPF and the chosentopology must be such that the worst case rejection specification ismet. Because Q_(l)=f₀/BW, as the 3 dB bandwidth (BW) decreases, the I.L.increases. Thus, if BW is too small relative to f₀, the resulting BPFwill have an unacceptably high I.L., requiring a tradeoff between BW andI.L. For practical designs, a low-order tunable BPF should have thelowest possible I.L. consistent with meeting the worst-case requiredrejection. Some topologies are preferred in that they naturally providea low side (below the transmission band) zero or a high side (above thetransmission band) zero.

[0053] A topology such as that shown in FIG. 8a, where the resonators404 and 408 are electromagnetically coupled along their entire length,is one such topology. It produces a high or low side zero depending uponthe capacitance of capacitor 432. This zero allows for a wider BW to beused, along with a BPF topology of lower ripple (resulting innumerically smaller g₁ values), thus providing a lower I.L.₀ as seen inequation (1).

[0054] Second, the low-order tunable filter must be tunable to cover theentire BW, just as with a fixed-tuned filter. Finally, the tunablecapacitor used within the low-order tunable filter should be ofsufficiently low loss so the resulting filter has an I.L. that meets orexceeds specifications. Although a tunable 1^(st) or 2^(nd) orderbandpass filter will be of minimum added loss compared to a higher order(N>2) fixed-tuned bandpass filter design, the tunable component(variable f-e capacitor) must have a fast tuning mechanism and betunable to cover the entire bandpass range, using the available tuningvoltage.

[0055] The total loss of a capacitor, L_(t), whether tunable or not, isgiven by a ratio of its dissipated to stored energy, where the energy isstored in the electric field and dissipated in resistance, i.e.,L_(t)=(dissipated energy)/(stored energy). The inverse of this loss isthe quality factor, Q=1/L_(t).

[0056] For a capacitor, L_(t) may be given by the quantity (ω*R_(s)*C),where ω is the frequency in radians, R_(s) is the total seriesresistance of the capacitor, and C is the capacitance. This definitionof Q is valid for frequencies which are below self resonance due to thereactance of stray inductance associated with any real capacitor andabove the frequency at which Rp effectively shunts C, that is, thecapacitive reactance is much smaller than Rp. Over this range offrequencies, a real capacitor can be modeled as a resistance Rs inseries with the desired capacitance. Thus, Q is inversely proportionalto the quantity (ω*R_(s)*C). For example, as any of the quantities, C, ωand R_(s) is increased or decreased, all other factors being heldconstant, Q decreases or increases, respectively. Or, to keep Qconstant, if one of the quantities, C, ω, or R_(s) is decreased orincreased the product of the other two quantities must be increased ordecreased, respectively.

[0057] The importance of determining the total loss given by an f-ecapacitor in a resonant circuit can be seen from the followingequations: L_(c)=1/Q_(c) and 1/Q_(T)=1/Q_(c)+1/Q_(u), where,

[0058] Lc=the loss of the capacitor;

[0059] Q_(T)=the total Q of the f-e capacitor and the resonator orinductor combined;

[0060] Q_(c)=the Q of the capacitor; and

[0061] Q_(u)=the Q of the unloaded resonator or alternatively, the Q ofan inductor used to create a parallel resonant circuit.

[0062] As Q_(c) increases, it will affect the Q_(T) less and less. IfQ_(c) is infinite, it has no affect on Q_(T). For practical purposes,this is also true if Q_(c) is approximately 10*Q_(u). The converse istrue too. As Q_(u) becomes higher and higher relative to Q_(c), Q_(u)has less and less effect on Q_(T). In either case, the highest practicalQ_(c) is desired.

[0063] For example, in the PCS band, for a 1.0 pF tunable capacitor tohave a Q_(c)=250 at 2.0 GHz requires that R_(s) be 0.32 Ω (ohms). Thisassumes Rp, the parallel resistance, which shunts C, is much greaterthan Zc, the impedance of the capacitor, at 2.0 GHz (where Rp>about 1.6kΩ here, for example, as the absolute value of Zc=0.0126 Ω), and thatthe capacitor self resonant frequency is well above 2.0 GHz, so that theseries inductance is negligible. To minimize loss (obtain a low R_(s))requires an accounting of all loss mechanisms present and an eliminationof these loss mechanisms if possible.

[0064] For f-e devices, the total loss is governed by summing eachsource contribution as follows:

L _(t) =L _(geom) +L _(attach) +L _(metal) +L _(sub) +L _(rad) +L_(meas) +L _(f-e);

[0065] where L_(geom) is derived from the topology of the capacitor,

[0066] L_(attach) is loss due to device attachment,

[0067] L_(metal) is the total metal loss,

[0068] L_(sub) is the base substrate loss (if present),

[0069] L_(rad) is the radiation loss, both desired and undesired,

[0070] L_(meas) is the total loss arising from measurement errors, and

[0071] L_(f-e) is the f-e loss tangent.

[0072] This loss allocation can first be used to obtain an accuratevalue of L_(f-e) (or f-e tan δ) at the desired operating frequency inthe manner in which the f-e capacitor will be used. To correctly deriveL_(f-e), one must eliminate or constrain all of the other losscontribution sources just described. For example, L_(geom) will varyaccording to topology, being best for an overlay capacitor, worse for agap capacitor, and much worse for an IDC capacitor. Although this losscan be reduced and controlled, it is inherent to a device. Consequently,the choice of topology for a given f-e capacitor will affect the bestpossible Q_(c) attainable from the f-e capacitor. Electromagnetic (EM)software can establish a baseline loss for a desired geometry, assuminga lossless f-e film. This baseline loss represents the best (lowest)loss for a given geometry.

[0073] In general, a gap capacitor is easiest to fabricate. An IDC isnext easiest, and an overlay capacitor is hardest of these three.Compared to an IDC, the gap capacitor will have a better Q but lowercapacitance per unit cross section (W in FIG. 1a). The IDC's capacitanceis greater due to the use of a number of fingers per unit cross section.For many communication filter applications, however, large capacitance(C≧4.0 pF) is not needed. Thus, a gap capacitor often can provideadequate capacitance. The inherently high value of κ for most f-e filmshelps provide relatively high capacitance per unit cross section, W,compared to a conventional gap capacitor.

[0074] L_(attach) arises from discrete device attachment techniques,including, for example, solder, silver paint, or wire bonding. Theseattachment losses may be large and unpredictable. The lowest losses areachieved by direct fabrication of the f-e capacitor to the resonator orother RF circuitry, thus minimizing if not eliminating this losscomponent.

[0075] The inherent loss of a stand-alone f-e capacitor is of littleconsequence. What is of much greater consequence is any added lossarising from the attachment of the f-e capacitor to a circuit. Even ifthe f-e capacitor were lossless, should a large loss connection be used,the overall effect is that of a lossy f-e device. For example, if aQ≧250 at 2.0 GHz is desired for a capacitance of 1.0 pF, then the totalseries resistance R_(s) must be ≦0.32 ohm. Any additional loss will thusfurther reduce the Q of this capacitor. That this additional loss isexternal to the actual capacitor is irrelevant. Even unavoidable lossmechanisms, such as those due to mounting, for example, lower theeffective Q of the capacitor from the perspective of its effect on thesystem.

[0076] For minimum added loss, the connection between the f-e capacitorand the resonator should provide the lowest added resistance. Thus, theelectric currents and charges associated with the f-e capacitor shouldsee a minimum added loss. Conventional bonding or mounting techniques,such as (but not limited to) soldering, wire bonding or silver paint orpaste do not provide for such a low loss, controllable bond.

[0077] The added, unpredictable loss arising from the use of suchbonding methods degrade the realized Q regardless of whether or not thef-e capacitor is being used for resonator tuning purposes orcharacterization of an f-e film. Thus, for best performance (lowestloss) the f-e capacitor structure should be directly fabricated onto orwith the resonator it is meant to tune or onto other essential RFcircuitry. Only by direct fabrication can there be a minimum losstransition for electromagnetic (EM) sources (currents) from the f-etuning elements to the resonator. The desirable effects of direct f-ecapacitor fabrication onto or with a resonator can be enhanced by thelack of sharp corners or transitions.

[0078] Factors for L_(metal) include the surface roughness (SR) of themetal, metal thickness as compared to skin depth, δs, and conductivity.SR may be effectively eliminated as a factor if SR is less thanaproximately 10 micro inches root mean square (rms) for operatingfrequencies in the L and S band (1-4 GHz). The metal thickness may bereduced as a factor if the thickness is 1.5δs or greater, or effectivelyeliminated if the thickness is ≧5δs. For electrode contacts, metalthickness (t_(m)) can be approximately 1.5δs. For the case ofelectromagnetic resonators, where a travelling or standing wave must besupported, i.e., where the metal in question extends for an appreciablefraction of a wavelength (about 10% or greater), the metal thicknessshould be closer to about 5δs or greater.

[0079] Conductivity is best for silver, copper and gold (Ag, Cu, and Au,respectively). Thus, L_(metal) can be reduced and controlled, but noteliminated as a factor. Its effect, however, can be calculated byexpressions well known to those skilled in the art, or by using linecalculator tools available in commonly used circuit simulators, such asEagleware or Touchstone. Further, precise fabrication control can boundgeometric variations in L_(metal).

[0080] The loss contribution represented by L_(sub) may be minimized bychoosing a low loss substrate with a loss tangent less than 0.001 andpreferably less than 0.0005 at the operating frequency of interest.Suitable materials include >99% pure alumina, a best current choice forloss/cost benefits. Sapphire or MgO are better than alumina in that theyhave lower loss tangents, but they are more expensive. All thesematerials will accept f-e thin films without buffer layers and have asurface roughness that is acceptable with little or no furtherpolishing. Semiconductor substrates are poor choices because of theirrelatively high conductivity. In addition to the factors of losstangent, surface roughness and price, suitable substrates should not bebrittle, can be fabricated as larger area wafers, and can be easilymetallized without extensive preprocessing.

[0081] Separating out L_(sub) from the total loss of a compositesubstrate (f-e film plus substrate) can be achieved by using EM field orcircuit simulation software. For example, Sonnet, Momentum, or IE3D maybe used. Thus, L_(sub) can be reduced significantly and calculatedprecisely.

[0082] L_(rad) can be eliminated by proper shielding and design, and sois typically not a factor. It should be noted that a wide variety offilters, especially planar filters such as combline or hairpin, dependupon radiative coupling to achieve their desired performance. In thesecases, one should ensure that the unwanted, stray coupling is reduced,if not eliminated.

[0083] L_(meas) can add significantly to the circuit loss error becausesmall, added loss significantly reduces the measured Q of thedevice-under-test (DUT) or system thus obscuring the intrinsic Q of theDUT. The conventional method for measuring dielectric constant and losstangent in a material is the cavity perturbation technique, which iswell known to anyone skilled in the art. At L-band, however, the size ofthe cavity becomes quite large. When characterizing thin films (asopposed to bulk) with film thickness ≧1.5 μm, such as f-e films, theproblem becomes very difficult as measurement errors can be severe.Furthermore, one should characterize an f-e capacitor (or filter) in amanner most similar to how it will be used. Thus, the preferred way tocharacterize f-e compounds or films is by microstrip resonatortechniques.

[0084] For the purposes of determining f-e film characteristics andcharacterizing f-e capaictors, microstrip techniques are preferred to,for example, stripline or other volumetric techniques for f-e filmcharacterization for the following reasons:

[0085] 1) Microstrip circuits are planar systems with no substrate as atop cover (which would be a stripline circuit), so no bonding of hardsubstrates as top covers is required. So there is also no need forcontinuity of ground planes (top to bottom) as needed in a stripline,for example.

[0086] 2) Preferably gap capacitors, and alternatively, IDC's, can bereadily fabricated and measured.

[0087] 3) A large body of knowledge exists as to the characterization ofmicrostrip resonators.

[0088] 4) No complex fixturing or fabrication or both are needed as arerequired for dielectric cavities, for example.

[0089] One should measure high-Q circuits using resonator techniquesbecause broadband measurement may not accurately resolve sub-ohmresistive losses at RF/microwave frequencies with any accuracy. For thesame reason, LRC meters are not a good choice.

[0090] Measurement at radio frequency is requried to correctly obtain Q,and consequently Rs, for an f-e capacitor, since low frequencymeasurement, especially those below about 10 to 100 MHz, is dominated bya large parallel resistance, Rp, that shunts the capacitance inquestion. The dominance of Rp, along with the relatively small values ofthe capacitance in question (≦4.0 to 5.0 pF) prevents reliable Q (andtherefore Rs) measurement at low frequencies.

[0091] When used to measure losses, wafer probe stations must becarefully used because it is difficult to calibrate out resistive andinductive loss at RF/microwave frequencies. Probe tips along with theirground connections are also sensitive to placement on the DUT as well asthe pressure used to apply them. As a consequence, it is better to use aresonant test circuit that allows for direct measurement of the desiredparameters in a way that does not require individual device lossmeasurements.

[0092] Thus, for measurements on resonant circuits, a network analyzeris the preferred choice. To minimize measurement loss and attain themost accurate measurement, one should calibrate out loss to the DUT,perform a full two port calibration of the network analyzer, and useaveraging for calibration and measurement. Finally, proper analysis ofthe measured data, such as that outlined in “Data Reduction Method for QMeasurements of Strip-Line Resonators,” IEEE Transactions in MTT, S.Toncich and R. E. Collin, Vol. 40, No. 9, September 1992, pp. 1833-1836,hereby incorporated by reference, is required to accurately extract theQ, or loss, of the capacitor under test.

[0093] Using the results of above discussion to minimize, eliminate, orbound each of the foregoing losses, the total loss may be re-expressedas:

L _(t) =L _(geom) +L _(metal) +L _(f-e) +ΔL _(misc)

[0094] As discussed above, both L_(geom) and L_(metal) may be quantifiedand removed analytically to obtain an accurate measure of L_(f-e).L_(geom) can be determined from an accurate electromagnetic simulationof the circuit based on a lossless f-e material assumption. L_(metal)can be determined using the expressions for metal loss assumingconductivity, SR (if applicable), and skin depth. The final term,ΔL_(misc), represents a combination of the incomplete removal of theother loss mechanisms or from the finite bounds on or incomplete removalof L_(metal) and L_(geom) or both. As such it represents an irreducibleerror term. For accurate measurements of f-e film/component properties,it should be minimized and bounded, as described in the precedingsections.

[0095] Finally, to reduce the effect of L_(f-e) to a minimum one mustuse selective f-e film deposition to place the f-e film only in regionswhere it is needed for tuning and nowhere else.

[0096] The process of accounting for all loss mechanisms and eliminatingor bounding these losses not only determines f-e loss but alsoestablishes correct design guidelines for low-loss tunable filters.Knowledge of L_(f-e) gives the designer a baseline for the f-e film thatis necessary for doing any type of optimum design using f-e films. Thisknowledge is necessary if one is to effectively trade-off loss tangentfor tunability, for example. In short, accurate fabrication andmeasurement techniques result in consistent f-e film losscharacterization and application.

[0097] Given the above techniques for minimizing loss, preferredembodiments for the three types of f-e capacitors may now be discussed.It will be appreciated, that although these designs are for use in the Lband (1-2 GHz), the teachings of the present invention may be used todesign f-e capacitors for other frequency bands.

[0098] A preferred f-e tunable gap capacitor 10 is shown in FIGS. 1a and1 b for use in the cellular band (800 to 1000 MHz) and the L-band (1-2GHz) for wireless handsets. The gap capacitor 10 is preferably formed ona ≧99% pure, 0.5 to 1.0 mm thick alumina, MgO, or sapphire substrate 12,having an SR less than a 5.0 micro inch RMS. Alternatively, the gapcapacitor can be directly patterned on the front or rear face or aside-wall of any number of resonators structures. Examples are coaxial,monoblock or stripline resonators. Such a capacitor should be fabricatedas close to its point of electrical connection to the resonator aspossible.

[0099] The substrate 12 may have a metal ground plane 14 depending onother requirements. However, the preferred embodiment is without aground plane to minimize stray capacitance. Preferably, a f-e layer 16of approximately 0.1 to 2.0 microns in thickness formed of BSTO or othersuitable or desirable f-e material for maximum capacitance and tuningrange is deposited on the substrate 12. More preferably, layer 16 is 0.5to 1.0 microns in thickness. The Ba/Sr fraction, doping, alloying,mixing with other components, and/or annealing determine the desiredtuning characteristics and loss (tan δ), and therefore Q also.

[0100] Generally, it is preferred that the tuning characteristics meetthe minimum required tuning range with the minimum tuning voltage.Preferably, x=0.5 in the Ba_(x)Sr_(1−x)TiO₃ composition for roomtemperature operation, regardless of doping with other elements and pre-or post-process annealing. It will be appreciated that other f-ematerials beside BSTO may be used as well. A metal layer 18 formed onthe f-e layer 16 defines a gap 20 that is preferentially 3.0 to 5.0microns wide. Preferably, metal layer 18 is 0.5 to 6.0 microns thick.More preferably, metal layer 18 is 1.5 to 2.5 microns thick. It will beappreciated that the gap 20 can be wider or narrower than this rangedepending on requirements and processing equipment. For minimum addedloss in the PCS band, the resulting capacitance will be approximately0.6 pF to 1.5 pF at 0 volts DC while for the cellular CDMA band it willbe about 1.0 pF to 3.0 pF. The width of the capacitor, W 17, willfurther determine the f-e capacitance, depending on the particular f-efilm used and the desired gap 20. The width will typically be from 0.25mm to 2.0 mm. The capacitance is typically 0.6 to 3.0 pF. The resultingcapacitor should provide a Q of at least 80 at 2.0 GHz to meet theexisting worst case CDMA PCS band BPF loss specification.

[0101] To minimize the added loss from the f-e film, selectivedeposition must be used, i.e., the f-e film is deposited only whereneeded for tuning and nowhere else as stated above. For example, in thegap capacitor 20 of FIG. 1a, one could deposit the desired f-e film 16in a narrow region D_(f-e) around the gap 20, as shown in FIG. 1a.D_(f-e) should be large enough to ensure that the gap 20 can berepeatedly patterned over the f-e film in manufacturing (allowing formask alignment tolerance) and to cover the needed area under the gap 20for tuning purposes. For the L-band PCS filters, D_(f-e)=0.2 to 0.5 mmis adequate with 0.2 mm preferred. As the operating frequency increasesD_(f-e) can decrease. As the operating frequency decreases, D_(f-e) canincrease.

[0102] F-E film properties and fabrication will play a significant rolein overall capacitor loss. Many techniques exist to mitigate andminimize f-e film loss. One feature of f-e films is that f-e film lossand tunability usually have an inverse relationship. That is, theyusually must be traded off against each other. The greater the f-e κtuning range, the greater the f-e loss in most cases.

[0103] Thus, even though f-e materials can achieve a κ tuning range ofabout 3 to 1, less tuning may be acceptable for a given filterapplication. In that case, less tuning would be chosen, with the benefitof less loss. For example, in the U.S. PCS CDMA band, the tuningrequirement in the transmit band is from 1850 MHz to 1910 MHz, or about4%. Thus, the f-e material can have significantly less tunability than 3to 1.

[0104] For example, an f-e gap capacitor with 0.6 pF at 0V DC bias,needs to tune 33%, (from 0.6 pF down to 0.4 pF) to tune over the PCStransmit band. The actual tuning range depends on the BPF topology andthe band over which the BPF must be tuned. The required tuning voltageto provide the 33% tuning in this example depends on the f-e capacitorgeometry, including f-e film thickness, and the f-e filmcharacteristics.

[0105] The effect of κ tunability on frequency tunability is determinedby the filter topology. This effect must also be considered in choosingan f-e material. But without accurate characterization of the f-e lossto f-e κ tunability trade-off, a designer cannot even begin to choose anoptimum f-e material. Accurate characterization of this trade-off allowsa designer to choose an optimum f-e material (providing the lowest losswhile meeting the tuning requirements).

[0106] With respect to L_(geom) for a gap capacitor, the majorcontributions to loss are the four corners formed by the gap. Theselosses can be reduced by rounding the corners.

[0107] In comparison to gap and interdigital capacitors, an overlaycapacitor has the lowest L_(geom). An overlay capacitor is an example ofa parallel plate geometry where the plate dimensions (length and width)are much greater than the plate separation. Given such a geometry, mostof the electric field between the plates is uniform except for fringingalong the edges. The fringing effect can be reduced significantly by theuse of a guard band, as is well known in the art. Thus, the geometricloss from a parallel plate capacitor is quite low. In addition, parallelplate geometries can provide high capacitances along with high tuningfrom small control voltage swings.

[0108] A preferred overlay capacitor 30 is illustrated in FIGS. 2a, 2 b,2 c, and 3 that minimizes contributions to L_(geom). The capacitor 30 isdeposited directly on a 25 mil alumina substrate 31. A first metal layer34 bonds to the substrate 31. The shape of metal layer 34 is alsoillustrated in FIG. 2b. A ferro-electric layer 36 overlies the metallayer 34. To form the overlay capacitor 30, a metal pad 40 formed on theferro-electric layer 36 overlaps a portion of the first metal layer 34.FIG. 3 illustrates an enlarged view of the overlapping portions. Boththe metal pad 40 and the metal layer 34 have a tapering region thatforms an overlay capacitor 30 of the appropriate capacitance. Anadditional metal pad 41 overlaps the metal layer 34 to form a DCblocking capacitor 42. The metal pad 41 is tapered to form anappropriate capacitance for the DC blocking capacitor 42.

[0109] Due to the high dielectric constant (κ) of the most likely f-efilms to be used, the overlay capacitor 30 may be quite small in areaand still provide a capacitance (C_(f-e)) of 1.5 pF. A bonding bias pad44 is provided for attachment of a high value (500-1000 kΩ) chipresistor. Note that the f-e film is deposited not only under the overlaycapacitor 30 but also the blocking capacitor 42. However, the effect onthe capacitance (C_(DC)) of the DC blocking capacitor 42 is irrelevantif C_(DC)≧180 pF and C_(f-e)≦1.5 pF, even under maximum V_(DC) bias(preferably 10V DC). This is because the DC blocking capacitor has ahigh enough capacitance that even when the capacitance is reduced by f-etuning, it still has a minimal effect on C_(f-e).

[0110] In such an embodiment, 0.7≦C_(f-e)≦1.5 pF, f-e κ is approximately1000, the overlapped portion of the metal pad 40 forming the overlapcapacitor 30 is approximately 7.0 μm×7.0 μm, and the f-e film thicknessis approximately 1.0 μm. The metal layer 34 may be Pt and have athickness of ≦0.5 μm. The metal pads 40 and 41 may be Ag and have athickness of approximately 1.5-2.5 μm.

[0111] While the L_(geom) of an overlay capacitor is lower than that ofa gap capacitor, L_(f-e) of an overlay capacitor may be higher, as allof the rf field is concentrated in the f-e film. In a gap capacitor therf field is partially in air, partially in the f-e film and partially inthe substrate. For the same reasons, an overlay capacitor has greatercapacitance tunability for a given applied voltage than a gap capacitor.

[0112] For a given cross sectional area, an IDC can provide a highercapacitance than a gap capacitor. It is more lossy, however, with themain contributions to L_(geom) including the gap spacing; loss increasesas the gap spacing decreases. Similarly, loss increases as finger widthdecreases. The finger length also affects loss with loss increasing asfinger length increases; especially in a microstrip (the most common)realization of an IDC as the odd mode loss dominates in such astructure. In addition, loss increases as the number of fingersincreases due to loss introduced from the additional sharp corners; notethat increasing the number of fingers is typically used to increase thecapacitance of an IDC.

[0113] Many investigators in the f-e area have used IDC's with narrowfinger widths and gaps (≦5.0 μm for each) to characterize f-e film. Thisis problematic, as such an IDC structure gives a high L_(geom) andtherefore a low Q by itself. Typically, Q is much less than 100 at 2.0GHz for about 1.0 pF, even without any L_(f-e). This makes it quitedifficult to measure L_(f-e). The wide spread use of broad bandmeasurement techniques, as described above, further obfuscates anyL_(f-e) measurement.

[0114] A preferred IDC capacitor 60 is illustrated in FIG. 4 thatminimizes the contributions to L_(geom). It is formed on a 99.5% purealumina, MgO, sapphire or other suitable substrate 62 of thickness ofapproximately 0.2 to 1.5 mm. A f-e layer 64 is formed on the substrate62. An input port 66 and output port 68 couple to the IDC capacitor 60.A metal layer 70 having a thickness of 1.5 to 3.0 microns and depositedon the f-e layer 64 forms a gap spacing 72 of approximately 5.0 micronsand a finger width 70 of about 150 microns, or greater if possible.

[0115] A general methodology for constructing a tunable bandpass filtermay now be described. As a first step, a designer must tradeoff the 3 dBbandwidth of the tunable filter with filter order to achieve therequired out-of-band rejection. As is well known, as the filter order isincreased, its rolloff rate increases, making it easier to achieve arequired rejection specification. The rolloff is modeled as beginning ateither of the 3 dB points defining the 3 dB bandwidth (BW). Thus, as theBW is decreased, it also becomes easier to achieve a required rejectionspecification.

[0116] For minimum loss the lowest order filter is desired. Typically,this will be a 2^(nd) order BPF. A low order BPF has a further advantageof being simpler to fabricate and tune, using fewer tunable resonators.

[0117] A Chebychev prototype BPF is preferred over a Butterworth as thisgives the designer flexibility to trade off passband ripple without-of-band rejection. The designer should strive to meet the worst caserejection specification by bandwidth adjustment without the addition ofextra transmission zeros as transmission zeros increase filtercomplexity, cost and loss at the corresponding passband edge. One can,however, exploit topologies that have naturally occurring high or lowside transmission zeros in this case.

[0118] Narrowing BW too much, however, will increase the insertion loss,as discussed above. Thus, the narrowest BW should be chosen that meetsthe required rejection specification over all specified operatingconditions. If the chosen BW provides an unacceptable insertion loss,the BW should be increased, perhaps also requiring an increase in filterorder or increased passband ripple (if acceptable). An additional highor low side transmission zero may be added if desired.

[0119] A tunable BPF requires control circuitry. This is an addedexpense, not required of fixed-tuned BPF's. Thus, a desirable tunablefilter design should provide a decreased insertion loss, smaller size,or other benefit over that of a fixed-tuned BPF while meeting rejectionspecifications, to offset this expense. To achieve decreased insertionloss and smaller size, it is preferable to use no more than a one or twostage tunable filter. However, it will be appreciated that theprinciples of the invention may be advantageously used to design tunablef-e filters of arbitrary order.

[0120] Given a choice for filter order and BW that satisfies therejection requirements, the highest possible Q_(u) for a resonatorshould be used to meet or exceed the required I.L., given size andheight constraints. To define Q_(u), a topology should be chosen for thebasic stage 100 illustrated in FIG. 5. Each stage 100 is formed by aresonator 102 coupled to a f-e capacitor 104. The f-e capacitor 104 mayassume one of the forms described herein. The resonator 102 is shown asa grounded quarter wavelength resonator but an open circuit one-halfwavelength resonator may also be used. Moreover, the resonator may be ofother suitable electrical length.

[0121] The basic stage 100 may be considered a tunable EM resonator. Thef-e capacitor 104 may be coupled either in series or in shunt with theresonator 102 as determined by the nature of their connection. As shownin FIG. 6, the f-e capacitor 104 is coupled in shunt with the resonator102 such that Q_(f-e) of the f-e capacitor 104 affects the Q of thefix-tuned EM resonator 102. Volumetric resonators (e.g., coaxial,stripline, and monoblock) are preferred, as they provide the highestQ_(u) and smallest area and height at a minimal price compared toplanar, i.e., microstrip or coplanar waveguide (CPW) alternatives.

[0122] Whether a tunable capacitor is placed in series or shunt with avolumetric resonator is often determined by the case of the connection;sometimes only one placement is even possible. Another key feature indetermining whether a tunable capacitor is placed in series or shunt isthat of minimum added loss and to a lesser extent, tuning range. A shuntconnection will typically produce a more physically compact tunablefilter than a series connection. It is usually easier to achieve aswell. A series connection can provide better tuning in the case whereelectromagnetically coupled resonators (like monoblock, coaxial orstripline) are coupled along their entire extent (rather than coupledthrough a small aperture). A series connection is a more natural choicein these cases from a fabrication perspective.

[0123] As discussed above, the attachment losses may be significant ifthe f-e capacitor 104 is not integrated with the resonator 102 or otherRF circuitry. Once a topology for the f-e capacitor 104 is chosen, itsQ_(c) may be derived as discussed above. Q_(T) for the overall basicstage 100 is then given by 1/Q_(T)=1/Q_(c)+1/Q_(u),

[0124] where Q_(u) is the unloaded Q of the resonator 102;

[0125] and Q_(c) is the Q of the f-e capacitor.

[0126] Given the Q_(T) for the basic stage 100, the designer may useequation (1) to determine if the required I.L. will be achieved orexceeded. Should the I.L. be too high, the designer may obtain a lowerI.L. by increasing either or both of Q_(c) or Q_(u). If either Q_(c) orQ_(u) cannot be increased further, they will ultimately limit Q_(T).Further reduction of I.L._(o) can then be obtained only by switching toa lower loss topology. For example, Q_(u) may be increased if avolumetric instead of a microstrip resonator is used for a givenfootprint (area).

[0127] For high volume applications, such as CDMA wireless handsets,transverse electromagnetic (TEM) wave volumetric resonators arepreferred. Such volumetric resonators can be either ceramic loadedcoaxial resonators, slabline (monoblock) or stripline, to name the threemost common realizations. The standard narrow band (typically defined asa BW≦10% of f_(o)) topology can be realized using top capacitivelycoupled (TCC) BPF's fabricated with either coaxial or striplineresonators. The TCC topology as shown in FIG. 8, lends itself to shuntf-e tuning, as this provides the most compact realization (having asmaller footprint than a TCC topology with series f-e tuning). Sincegrounded quarter wave resonators behave as parallel LC tuned circuitsnear resonance, placing an f-e tuning capacitor in shunt isadvantageous.

[0128] Stepped impedance realizations of monoblock BPF's can be used aswell. Monoblock resonators are typically EM coupled along their entirelength, a direct consequence of their design. While they lend themselvesto series f-e tuning, shunt tuning can be effectively used as well.Their electrical lengths can be tuned by the selective deposition andpatterning of f-e tuning capacitors. Non-TEM resonators can be used aswell, including, but not limited to, dielectric loaded waveguideresonators or dielectric pucks (with or without a shielded enclosure).

[0129] However, height restrictions may limit the achievable Q_(u) fromvolumetric resonators. An alternative to a volumetric coaxial resonatorin such height-constrained systems is to use a stripline resonator.Here, one can make the center conductor wider (up to a point) thusimproving Q_(u) while keeping the total height fixed. This embodimenthas further merit in that the incorporation of a planar f-e capacitorsuch as a gap capacitor or IDC can be realized efficiently by making thetop cover of the stripline resonator end before the location of the f-ecapacitor. In this manner, the planar f-e capacitor would be formed onthe portion of the substrate forming the bottom cover of the striplineresonator that extends beyond the top cover.

[0130] The formation of a “pedestal” on which the f-e capacitors can beoptimally integrated as shown in FIG. 11a for a TCC structure usingcoaxial resonators as an example. The f-e capacitors are integrated asextensions of the input and output capacitors 315 a and 315 b in FIG.11a on the pedestal. Alternatively, the f-e capacitors can be patternedand fabricated on the open ends (faces) (not shown) of the coaxial ormonoblock resonators.

[0131] Regardless of the particular resonator being implemented, ifheight restrictions prevent any further increase in the Q_(u) of theresonator, Q_(c) would have to be increased instead by, for example,replacing an IDC f-e capacitor with a gap or an overlay f-e capacitor.

[0132] For many applications a single stage bandpass filter 140 will besatisfactory as illustrated in FIG. 6. As discussed with respect to FIG.5, the bandpass filter 140 will include the f-e capacitor 104 and theresonator 102. A variable DC voltage 142 applied to the f-e capacitor104 tunes the filter 140. The RF signal to be filtered is applied atinput port 144 and is output at output port 146. Note that the inputport 144 and output port 46 are interchangeable. A capacitor 143 isdefined both between the input port 144 and the resonator 102. Anothercapacitor 145 is defined between the output port 146 and the resonator102. The f-e capacitor 104, regardless of whether it is a gap, overlay,or IDC capacitor, is constructed to minimize losses in the mannerdescribed above. Similarly, the resonator 102, which may be either ashorted ¼ wavelength resonator or a ½ wavelength open circuit resonator,is selected to maximize Q_(u).

[0133] A higher Q_(u) will be provided by a volumetric resonator such asa coaxial resonator, a dielectric loaded waveguide, a monoblock, or astripline resonator in a smaller footprint and at a lower cost.Alternatively, a larger area planar resonator such as a microstripresonator may be used if specifications and price constraints permit.Most microstrip resonator circuits would be fabricated by thin filmprocess on a hard substrate. As a result, they achieve less metalthickness than TEM resonators like coaxial and monoblock resonators thatare metalized by thick film processes. Microstrip resonators are oflarger size since part of the EM field is the air region above themicrostrip.

[0134] Turning now to FIG. 7, a planar realization 150 of the bandpassfilter 140 is illustrated. Resonator 102 is formed by a microstrip line152 grounded through via 154. Note that microstrip line 152 could alsobe terminated in a suitable lossless ground plane (not illustrated),obviating the need for via 154. Capacitors 153 and 155 are formed bygaps between the input and output microstrip lines 156 and 158 and theresonator microstrip line 152. It is desirable to make the capacitanceof capacitors 155 and 157 as large as practical (approximately 0.25 pF)to maximize input and output coupling while still maintaining a planarstructure. The microstrip lines are formed on substrate 157 of 99.5%pure alumina, MgO, or sapphire that is preferably of thicknessapproximately 1.0 mm for providing a maximum microstrip resonator Q. Thef-e capacitor 104 is formed as a gap capacitor by pad 160 andmircrostrip line 152, with f-e layer 162 underneath pad 160 andmicrostrip line 152.

[0135] A variable DC voltage source biases pad 160 through resistor 164.A DC blocking capacitor is positioned between pads 160 and 166, wherepad 166 includes a via 168 to ground. Note that pad 166 could also beterminated in a suitable lossless ground plane (not illustrated),obviating the need for via 168.

[0136] The DC blocking capacitor is needed if the resonator is shuntedas shown in FIG. 7. The capacitance of the DC blocking capacitor isideally at least 100C_(f-e) to minimize its loading effects on C_(f-e).Its Q is ideally ≧40 in the band of interest. It will be appreciatedthat the choice of a gap capacitor and a microstrip resonator isarbitrary—any of the forms discussed herein could be employed consistentwith the teachings of the present invention.

[0137] The bandpass filter of FIG. 7 may be ideally used as a testcircuit to characterize an f-e film as described herein. As such, thebandpass filter of FIG. 7 provides the following advantages:

[0138] 1) The f-e capacitor can be fabricated as it is to be used,particularly if that realization is a gap capacitor or IDC. Selectivef-e deposition is used.

[0139] 2) While an f-e gap capacitor is shown, an IDC could be used aswell. A gap capacitor has a simpler geometry. It is easier to fabricateand has lower geometric loss compared to an IDC. It is also easier tofabricate than an overlay capacitor.

[0140] 3) Since the circuit is fabricated with thin film processingtechniques the geometry can be precisely controlled and measured.

[0141] 4) Metal thickness can be accurately measured by profilometry.Metal type can be selected as desired (Au, Ag or Cu).

[0142] 5) A high Q microstrip circuit completes the fixed resonator partof the circuit.

[0143] 6) The f-e capacitor is directly fabricated in the resonator.There is no added loss due to soldering, bonding, etc. The transitionfrom resonator to f-e capacitor is uniform, or it can be tapered, ifdesired.

[0144] 7) No via holes are needed if large area ground planes and aWiltron test fixture (with jaws to hold and ground the circuit top andbottom) is used. Drilling vias in hard substrates is a significant costadder and reduces the number of such test circuits that can befabricated.

[0145] 8) This circuit can be accurately modeled in EM software.

[0146] 9) This circuit can be fabricated without f-e film to determine abase loss (at a higher f_(o), of course) of the circuit for correlationto simulations.

[0147] 10) The use of a low loss substrate minimizes its effect on theoverall circuit.

[0148] 11) Measured results of f_(o) and I.L_(o). can be used to extractf-e film dielectric constant and tan δ.

[0149]12) The circuit in FIG. 7 can be fabricated with an aperture inthe base substrate where the f-e cap is shown. Now, independent f-e capscan be placed over the aperture, held in place with pressure, allowingthe f-e caps to be tested as stand-alone components.

[0150] Referring now to FIG. 8a, a two stage TCC tunable BPF 400 isillustrated. As discussed with respect to FIG. 5, each stage of bandpassfilter 400 comprises a resonator 404 and 408 and f-e capacitor 410 a and410b. The resonators 404 and 408 are shown as ¼ wavelengthshort-circuited resonators but may also be ½ wavelength open circuitresonators. In either case, the resonator length is reduced by thepresence of C_(f-e).

[0151] A variable DC voltage applied to the f-e capacitors 410 a and 410b tunes the bandpass filter 400. The ferro-electric capacitors 410 a and410 b couple to ground through DC blocking capacitors 412 a and 412 b,since the resonators are shorted in this example.

[0152] An RF signal is received at input port 402 and output at outputport 406. Note that input port 402 and output port 406 areinterchangeable. In addition to input capacitor 434 a and outputcapacitor 434 b, which are functionally similar to capacitors 143 and145 discussed with respect to FIG. 6, an additional capacitor 432 isprovided as an impedance or admittance inverter between the resonators404 and 408 to create the desired BPF response. It will be appreciatedthat capacitor 432 can also be a discrete element or implemented throughaperture coupling between resonators 404 and 408.

[0153] The tunable two-stage filters 400 and 450 illustrated in FIGS. 8aand 8 b have a basic topology which creates a high or low side zero bythe addition of electromagnetic coupling along the entire length ofresonators 404 and 408. The zero can be used to provide better rejectionfor a given passband I.L. In the case of inter-resonator coupling alongtheir entire length, the passband I.L. and out-of-band rejection willchange as the ferro-electric capacitors tune the bandpass filter acrossthe passband. To minimize any resulting distortion, particularly in therejection band, capacitor 432 may be a f-e capacitor. Tuning capacitors413 and 419 makes the zero track in frequency with the tunable passband.

[0154] To facilitate the biasing and tuning of f-e capacitor couplingbetween the resonators 404 and 408, capacitor 432 may be replaced by f-ecapacitors 437 a and 437 b as shown in FIG. 8b. Capacitors 437 a and 437b ideally have a capacitance twice that of capacitor 432. In thisembodiment, the ferro-electric capacitors 410 a, 410 b, 437 a and 437 bmay all be tuned using a single DC tuning voltage VDC. Alternatively,different f-e materials can be deposited for capacitors 437 a and 437 bthan that used for capacitors 410 a and 410 b. Thus greater versatilitymay be obtained in tuning with a single voltage.

[0155] The single DC tuning voltage for the f-e capacitors may bearranged as shown in FIG. 9. In FIG. 9, V_(DC) is coupled to a dividernetwork 505. The divider network 505 is coupled to both f-e capacitors437 a and 437 b. The divider network 505 is configured to provide theappropriate tuning range to the f-e capacitors 437 a and 437 b so as tocause the zero to track with the passband, as described above.

[0156] The divider network 505 may be constructed as shown in FIG. 10.In FIG. 10, V_(Dc) is coupled to R_(1.) R₁ is coupled to R₂ and to bothcapacitors 437 a and 437 b. R₂ is also coupled to ground. R₁ and R₂ arechosen to cause the zero to track with the passband, as described above.

[0157] Alternatively, a separate voltage can be used to tune bothcapacitors 437 a and 437 b.

[0158] Turning now to FIG. 11a, a tunable two-stage filter 300 usingcoaxial, monoblock resonators 302 a and 302 b is illustrated. Note thatother resonator types could also be used. The resonators 302 a and 302 bmay be open or short circuited. The resonators 302 a and 302 b attach toa first surface of a substrate 301. Pads 304 a and 304 b formed on thefirst surface of the substrate 301 couple to the resonators 302 a and302 b through leads 305 a and 305 b. Pads 306 a and 306 b formed on thefirst surface of substrate 301 couple to pads 304 a and 304 b creatingthe desired gap for the ferro-electric capacitors 310 a and 310 b.Ferro-electric layers 312 a and 312 b underlying the pads 304 a and 304b and 306 a and 306 b complete ferro-electric gap capacitors 310 a and310 b. Note that the drawings are not to scale. Typically, the gapspacing is increased for clarity.

[0159] There are transmission lines 320 a and 320 b on a second surfaceof substrate 301. These transmission lines are used as input and outputports 320 a and 320 b for signals RF in and RF out. The input and outputcapacitors 315 a and 315 b are formed between the transmission lines 320a and 320 b and the pads 304 a and 304 b with substrate 301 in between,as shown in FIG. 11 b. FIG. 11b is a cross-sectional view of a portionof filter 300 shown in FIG. 11a. The cross-section is taken along lineB.

[0160] In addition, capacitor 321 is formed as a gap capacitor by theseparation of pads 304 a and 304 b. Note that the coupling provided bycapacitor 321 may alternatively be provided though aperture couplingbetween coaxial resonators 302 a and 302 b, eliminating the need forcapacitor 321. It will be appreciated that although the coaxialresonators 302 a and 302 b are shown as separate structures, they mayshare a common wall to save space and permit any aperture coupling.Additionally, there may be no space and no wall between them. I.e., theymay be mutually coupled monoblock resonators. In embodiments in whichthe coupling provided by capacitor 321 is implemented through aperturecoupling, the pads 304 a and 304 b would be separated by a sufficientdistance to minimize any gap capacitance between them. A bias voltageVDC couples through resistors 340 a and 340 b to tune the ferro-electriccapacitors 310 a and 310 b. Each of the ferro-electric gap capacitors310 a and 310 b couple to ground through DC blocking capacitors 341 aand 341 b.

[0161] Although the invention has been described with reference toparticular embodiments, the description is only an example of theinvention's application and should not be taken as a limitation.Consequently, various adaptations and combinations of features of theembodiments disclosed are within the scope of the invention asencompassed by the following claims.

I claim:
 1. A tunable electromagnetic signal filter comprising: adielectric constant adjustment signal generator for generating adielectric constant adjustment signal; a first element having acapacitance; a second element having an inductance; the first and secondelements configured as an electromagnetic signal filter having aresonant frequency; a ferro-electric material positioned proximate thefirst element for adjusting, responsive to the dielectric constantadjustment signal, the capacitance of the first element for adjustingthe resonant frequency; wherein a quality factor of the first element,when operated in a temperature range between about −50 degrees Celsiusand 100 degrees Celsius, is greater than about
 80. 2. The filter ofclaim 1, wherein the quality factor is greater than about
 180. 3. Thefilter of claim 1, wherein the quality factor is greater than about 350.4. A tunable electromagnetic signal filter comprising: a dielectricconstant adjustment signal generator for generating a dielectricconstant adjustment signal; an element having a capacitance and aquality factor; a volumetric resonator; the element and the resonatorconfigured as an electromagnetic signal filter having a resonantfrequency; a ferro-electric material positioned proximate the firstelement for adjusting, responsive to the dielectric constant adjustmentsignal, the capacitance of the first element for adjusting the resonantfrequency.
 5. The filter of claim 1 or 4, wherein the quality factor,when operated in a temperature range between about −50 degrees Celsiusand 100 degrees Celsius, is greater than about 80 in a frequency rangebetween 0.25 GHz and 7.0 GHz.
 6. The filter of claim 5, wherein thequality factor, when operated in a temperature range between about −50degrees Celsius and 100 degrees Celsius, is greater than about 80 in afrequency range between about 0.8 GHz and 7.0 GHz.
 7. The filter ofclaim 6, wherein the quality factor, when operated in a temperaturerange between about −50 degrees Celsius and 100 degrees Celsius, isgreater than about 80 in a frequency range between about 0.25 GHz and2.5 GHz.
 8. The filter of claim 7, wherein the quality factor, whenoperated in a temperature range between about −50 degrees Celsius and100 degrees Celsius, is greater than about 80 in a frequency rangebetween about 0.8 GHz and 2.5 GHz.
 9. The filter of claim 1 or 4,wherein the quality factor, when operated in a temperature range betweenabout −50 degrees Celsius and 100 degrees Celsius, is greater than about180 in a frequency range between 0.25 GHz and 7.0 GHz.
 10. The filter ofclaim 5, wherein the quality factor, when operated in a temperaturerange between about −50 degrees Celsius and 100 degrees Celsius, isgreater than about 180 in a frequency range between about 0.8 GHz and2.5 GHz.
 11. The filter of claim 1 or 4, wherein the quality factor,when operated in a temperature range between about −50 degrees Celsiusand 100 degrees Celsius, is greater than about 80 for a capacitance in arange between about 0.3 pF and 3.0 pF.
 12. The filter of claim 11,wherein the quality factor, when operated in a temperature range betweenabout −50 degrees Celsius and 100 degrees Celsius, is greater than about80 for a capacitance in a range between about 0.5 pF and 1.0 pF.
 13. Thefilter of claim 1 or 4 wherein the quality factor, when operated in atemperature range between about −50 degrees Celsius and 100 degreesCelsius, is greater than about 180 for a capacitance in a range betweenabout 0.3 pF and 3.0 pF.
 14. The filter of claim 13 wherein the qualityfactor, when operated in a temperature range between about −50 degreesCelsius and 100 degrees Celsius, is greater than about 180 for acapacitance in a range between about 0.5 pF and 1.0 pF.
 15. The filterof claim 1 or 4, wherein the resonator comprises a stripline resonator.16. The filter of claim 1 or 4, wherein the resonator comprises amonoblock resonator.
 17. The filter of claim 4, wherein the resonatorcomprises a coaxial dielectric loaded resonator.
 18. The filter of claim4, wherein a quality factor, when operated in a temperature rangebetween about −50 degrees Celsius and 100 degrees Celsius, is greaterthan about
 80. 19. The filter of claim 4, wherein a quality factor, whenoperated in a temperature range between about −50 degrees Celsius and100 degrees Celsius, is greater than about
 180. 20. The filter of claim1 or 4, further comprising a gap capacitor comprising: a substrate; ametal layer on a first surface of the substrate, the metal layerdefining the gap; a ferro-electric layer between the metal layer and thesubstrate; and wherein: the substrate is a low loss material having athickness between approximately 0.5 and 1.0 mm; the metal layer isformed to have a surface roughness less than approximately 5.0 microinch; and the gap spacing of the gap defined by the metal layer isapproximately 3.0 to 5.0 microns.
 21. The filter of claim 20, whereinthe low loss material is selected from the group consisting of magnesiumoxide, sapphire, and at least 99% pure alumina.
 22. The filter of claim20, wherein the metal layer has a thickness of approximately 1.5 to 2.5microns and the ferro-electric material comprises barium strontiumtitanate having a thickness of approximately 0.5 to 1.0 microns.
 23. Thefilter of claim 20, wherein the gap capacitor comprises metal lineshaving a width of approximately 0.25 mm to 2.0 mm.
 24. The filter ofclaim 20, wherein the ferro-electric material is localized in the areaof the gap.
 25. The filter of claim 24, wherein the ferro-electricmaterial is contained within about 2.0 microns of the gap.
 26. Thefilter of claim 20, wherein the metal lines defining the gap haverounded corners.
 27. The filter of claim 1 or 4, further comprising anoverlay capacitor comprising: a substrate; a first metal layer on afirst surface of the substrate, a second metal layer, and aferro-electric layer disposed between the first and second metal layers,wherein the ferro-electric layer has a thickness of approximately 1micron, the first metal layer has a thickness of less than approximately0.5 micron and the second metal layer has a thickness of betweenapproximately 1.5 to 2.5 microns.
 28. A method of designing a tunableferro-electric filter, the tunable ferro-electric filter constructed tooperate in a tunable frequency range, comprising: generating a filterdesign for the filter, the filter design having a tunable device;determining a minimum acceptable efficiency for the tunable filter, theminimum acceptable efficiency being represented by a quality factor, Q,of about 80 or more; selecting a tunable ferro-electric device for useas the tunable device in the filter design; confirming that the Q isgreater than about
 80. 29. The method of claim 28, wherein the Q isgreater than about
 200. 30. The method of claim 28, wherein the Q isgreater than about
 300. 31. A method of designing a tunable bandpassfilter, comprising: choosing a bandwidth and filter order to satisfy anout-of-band rejection and passband insertion loss requirement; choosinga topology for a ferro-electric capacitor; calculating thenon-ferro-electric losses for the chosen topology; choosing a resonatorhaving a first quality factor for coupling to the f-e capacitor; anddetermining, based upon the calculated non-ferro-electric losses andfirst quality factor, the required ferro-electric loss of theferro-electric capacitor to satisfy the insertion loss requirement forthe tunable filter.
 32. The method of claim 31, wherein the topologychosen for the f-e capacitor is a gap capacitor.
 33. The method of claim31, wherein the topology chosen for the f-e capacitor is an overlaycapacitor.
 34. The method of claim 31, wherein the topology chosen forthe f-e capacitor is an IDC capacitor.